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1 LT1533 ultralow noise 1a switching regulator features descriptio n u the lt ? 1533 is a new class of switching regulator designed to reduce conducted and radiated electromagnetic interfer- ence (emi). ultralow noise and emi are achieved by providing user control of the output switch slew rates. voltage and current slew rates can be independently programmed to optimize switcher harmonic content versus efficiency. the LT1533 can reduce high frequency harmonic power by as much as 40db with only minor losses in efficiency. the LT1533 utilizes a dual output switch current mode architecture optimized for low noise topologies. the ic includes two 1a power switches along with all necessary oscillator, control and protection circuitry. unique error amp circuitry can regulate both positive and negative voltages. the internal oscillator may be synchronized to an external clock for more accurate placement of switching harmonics. protection features include cycle by cycle current limit pro- tection, undervoltage lockout and thermal shutdown. low minimum supply voltage and low supply current during shutdown make the LT1533 well suited for portable applica- tions. the part may also be forced into a 50% duty cycle mode for unregulated applications. the LT1533 is available in the 16-pin narrow so package. n greatly reduced conducted and radiated emi (<100 m v p-p in typical application) n low switching harmonic content n independent control of switch voltage and current slew rates n two 1a current limited power switches n regulates positive and negative voltages n 20khz to 250khz oscillator frequency n easily synchronized to external clock n wide input voltage range: 2.7v to 23v n low shutdown current: 12 m a typical n easier layout than with conventional switchers n outputs can be forced to 50% duty cycle for unregulated applications n precision instrumentation systems n isolated supplies for industrial automation n medical instruments n wireless communications n single board data acquisition systems applicatio n s u typical applicatio n u , ltc and lt are registered trademarks of linear technology corporation. shdn duty sync col a col b pgnd r vsl r csl fb nfb LT1533 gnd v in 5v 14 2 15 16 13 12 7 11 3 4 5 6 10 820pf 16.9k 0.015 f 1000pf 15k note 1 33 f 15k 15k + t1 1n4148 b a l1 300 h l2 33 h 1n4148 2.49k 1% 8 9 21.5k, 1% c1 47 f 16v + c2 33 f 20v c1: sanyo os-con c2: avx tps tantalum l1: coiltronics ctx300-2 l2: coilcraft dt1608c-333 t1: coiltronics ctx02-13834 note 1: 25nh trace inductance or coilcraft b07t 12v 150ma 1533 ta01 + c t r t v c low noise 5v to 12v forward push-pull dc/dc converter b 2mv/div a 100 m v/div 2 m s/div 1533 ta02 12v output noise (bw = 100mhz) <100 m v p-p
2 LT1533 absolute m axi m u m ratings w ww u (note 1) input voltage (v in ) .................................................. 30v switch voltage (col a, col b) ............................... 30v shdn pin voltage .................................................... 30v feedback pin current ............................................ 10ma negative feedback pin current ............................ 10ma storage temperature range ................. C 65 c to 150 c maximum junction temperature ......................... 125 c operating junction temperature range LT1533c ............................................... 0 c to 100 c LT1533i ............................................ C 40 c to 100 c lead temperature (soldering, 10 sec).................. 300 c order part number LT1533cs LT1533is package/order i n for m atio n w u u electrical characteristics v in = 5v, v c = 0.9v, v fb = v ref . col a, col b, shdn, nfb, duty pins open, unless otherwise noted. consult factory for military grade parts. top view s package 16-lead narrow plastic so 1 2 3 4 5 6 7 8 16 15 14 13 12 11 10 9 nc col a duty sync c t r t fb nfb pgnd col b v in r vsl r csl shdn v c gnd t jmax = 125 c, q ja = 100 c/ w symbol parameter conditions min typ max units supply and protection v in recommended operating range l 2.7 23 v v in(min) minimum input voltage l 2.55 2.7 v i vin supply current 2.7v v in 23v, r vsl , r csl , r t = 17k l 12 18 ma i vin(off) shutdown supply current 2.7v v in 23v, v shdn = 0v l 12 30 m a v shdn shutdown threshold 2.7v v in 23v l 0.4 0.8 1.2 v i shdn shutdown input current C2 m a error amplifiers v ref reference voltage measured at feedback pin 1.235 1.250 1.265 v l 1.215 1.250 1.275 v i fb feedback input current v fb = v ref l 250 900 na fb reg reference voltage line regulation 2.7v v in 23v l 0.003 0.03 %/v v nfr negative feedback reference voltage measured at negative feedback pin with l C 2.550 C 2.500 C 2.420 v feedback pin open i nfr negative feedback input current v nfb = v nfr l C37 C25 m a nfb reg negative feedback reference voltage 2.7v v in 23v l 0.002 0.05 %/v line regulation
3 LT1533 electrical characteristics v in = 5v, v c = 0.9v, v fb = v ref . col a, col b, shdn, nfb, duty pins open, unless otherwise noted. symbol parameter conditions min typ max units g m error amplifier transconductance d i c = 25 m a 1100 1500 1900 m mho l 700 2300 m mho i esk error amplifier sink current v fb = v ref + 150mv, v c = 0.9v, v shdn = 1v l 120 200 350 m a i esrc error amplifier source current v fb = v ref C 150mv, v c = 0.9v, v shdn = 1v l 120 200 350 m a v clh error amplifier clamp voltage high clamp, v fb = 1v 1.33 v v cll error amplifier clamp voltage low clamp, v fb = 1.5v 0.1 v a v error amplifier voltage gain 180 250 v/v oscillator and sync f max maximum switch frequency 250 khz f sync synchronization frequency range f osc = 250khz l 375 khz r sync sync pin input resistance 40 k w v fbfs fb pin threshold for frequency shift 5% reduction from nominal 0.4 v output switches dc max maximum switch duty cycle duty pin open, r vsl = r csl = 4.9k, f osc = 25khz l 44 45.5 % duty pin grounded, forced 50% duty cycle 50.0 % t ibl switch current limit blanking time 200 ns bv output switch breakdown voltage 2.7v v in 23v l 25 30 v r on output switch-on resistance i col a or i col b = 0.75a l 0.5 0.85 w i lim(max) maximum current limit duty cycle = 15% 1 1.25 1.8 a short-circuit current limit duty cycle = 40% 0.8 a d i in / d i sw supply current increase during 16 ma/a switch-on time v dutyth duty pin threshold 0.35 slew control v slewr output voltage slew rising edge either a or b, r vsl , r csl = 17k 11 v/ m s v slewf output voltage slew falling edge either a or b, r vsl , r csl = 17k 14.5 v/ m s i slewr output current slew rising edge either a or b, r vsl , r csl = 17k 1.3 a/ m s i slewf output current slew falling edge either a or b, r vsl , r csl = 17k 1.3 a/ m s the l denotes specifications that apply over the full operating temperature range. note 1: absolute maximum ratings are those values beyond which the life of a device may be impaired. note 2: the LT1533 is designed to operate over the junction temperature range of C 4 0 c to 125 c, but is neither tested nor guaranteed beyond 0 c to 100 c for c grade or C 40 c to 100 c for i grade.
4 LT1533 typical perfor m a n ce characteristics uw switch current (a) 0 0.7 0.6 0.5 0.4 0.3 0.2 0.1 0 0.6 1533 g03 0.2 0.4 0.8 125 c 25 c 1.0 switch voltage (v) 85 c duty cycle (%) 0 ? i lim (ma) 0 50 100 150 200 250 300 10 20 30 125 c 25 c 40 1533 g02 50 change in i lim vs dc switch voltage drop temperature ( c) ?0 negative feedback voltage (v) nfb input current ( a) 150 1533 g05 0 50 100 2.30 2.35 2.40 2.45 2.50 2.55 2.60 2.65 2.70 35 30 25 20 15 ?5 25 75 125 v nfb i nfb negative feedback voltage and input current vs temperature feedback voltage and input current vs temperature switching frequency vs feedback pin voltage feedback pin voltage (v) 0 switching frequency (% typical) 0.1 0.2 0.3 0.4 1533 g07 0.5 120 100 80 60 40 20 0 0.6 error amplifier output current feedback pin voltage from nominal (mv) 400 ?00 ?00 ?00 error amplifier output ( a) 400 1533 g08 0 100 300 200 500 400 300 200 100 0 100 200 300 400 500 125 c ?0 c 25 c temperature ( c) v c pin voltage (v) 1533 g09 1.6 1.4 1.2 1.0 0.8 0.6 0.4 0.2 0 ?0 25 75 ?5 0 50 100 125 v c pin clamp voltage v c pin threshold v c pin threshold and clamp voltage vs temperature temperature ( c) ?0 feedback voltage (v) feedback input current ( m a) 1.30 1.29 1.28 1.27 1.26 1.25 1.24 1.23 1.22 1.21 1.20 2.0 1.8 1.6 1.4 1.2 1.0 0.8 0.6 0.4 0.2 0 0 50 75 1533 g04 ?5 25 v fb i fb 100 125 150 junction temperature ( c) input voltage (v) 2.60 2.65 2.70 1533 g01 2.55 2.50 2.45 ?0 0 50 75 ?5 25 100 125 150 minimum input voltage vs temperature temperature ( c) transconductance (mho) 1533 g06 2000 1900 1800 1700 1600 1500 1400 1300 1200 1100 1000 ?0 25 75 150 ?5 0 50 100 125 g m = ? i vc / ? v fb error amplifier transconductance vs temperature
5 LT1533 pi n fu n ctio n s uuu gnd (pin 9): signal ground. the internal error amplifier, negative feedback amplifier, oscillator, slew control cir- cuitry and the bandgap reference are referred to this ground. keep the connection to the feedback divider and v c compensation network free of large ground currents. v c (pin 10): the compensation pin is used for frequency compensation and current limiting. it is the output of the error amplifier and the input of the current comparator. loop frequency compensation can be performed with an rc network connected from the v c pin to ground. shdn (pin 11): the shutdown pin is used for disabling the switcher. grounding this pin will disable all internal cir- cuitry. normally this output can be tied high (to v in ) or may be left floating. r csl (pin 12): a resistor to ground sets the current slew rate for the collectors a and b. the minimum resistor value is 3.9k and the maximum value is 68k. current slew will be approximately: i slew(a/ m s) = 33/r csl(k w ) r vsl (pin 13): a resistor to ground sets the voltage slew rate for the collectors a and b. the minimum resistor value is 3.9k and the maximum value is 68k. voltage slew will be approximately: v slew(v/ m s) = 220/r vsl(k w ) v in (pin 14): input supply pin. bypass this pin with a 3 4.7 m f low esr capacitor. when v in is below 2.55v the part will go into undervoltage lockout where it will stop output switching and pull the v c pin low. pgnd (pin 16): power switch ground. this ground comes from the emitters of the power switches. in normal opera- tion this pin should have approximately 25nh inductance to ground. this can be done by trace inductance (approxi- mately 1") or with wire or a specific inductive component. this inductance ensures stability in the current slew control loop during turn-off. too much inductance (>50nh) may produce oscillation on the output voltage slew edges. col a, col b (pins 2, 15): these are the output collectors of the power switches. their emitters return to pgnd through a common sense resistor. col a and col b are alternately turned on out of phase. large currents flow into these pins so it is desirable to keep external trace lengths short to minimize radiation. the collectors can be tied together for simple boost applica- tions. duty (pin 3): tying the duty pin to ground will force the outputs to switch with a 50% duty cycle. the duty pin must float if not used. sync (pin 4): the sync pin can be used to synchronize the oscillator to an external clock (see oscillator sync in applications information section for more details). the sync pin may either be floated or tied to ground if not used. c t (pin 5): the oscillator capacitor pin is used in conjunc- tion with r t to set the oscillator frequency. for r t = 16.9k, c t(nf) = 129/f osc(khz) r t (pin 6): the oscillator resistor pin is used to set the charge and discharge currents of the oscillator capacitor. the nominal value is 16.9k. it is possible to adjust this resistance 25% to get a more accurate oscillator fre- quency. fb (pin 7): the feedback pin is used for positive voltage sensing and oscillator frequency shifting during start-up and short-circuit conditions. it is the inverting input to the error amplifier. the noninverting input of this amplifier connects internally to a 1.25v reference. this pin should be left open if not used. nfb (pin 8): the negative voltage feedback pin is used for sensing a negative output voltage. the pin is connected to the inverting input of the negative feedback amplifier through a 100k source resistor. the negative feedback amplifier provides a gain of C 0.5 to the feedback amplifier. the nominal regulation point would be C 2.5v on nfb. this pin should be left open if not used.
6 LT1533 operatio u in noise sensitive applications, switching regulators tend to be ruled out as a power supply option due to their propensity for generating unwanted noise. when switch- ing supplies are required due to efficiency or input/output voltage constraints, great pains must be taken to work around the noise generated by a typical supply. these steps may include precise synchronization of the power supply oscillator to an external clock, synchronizing the rest of the circuit to the power supply oscillator, or halting power supply switching during noise sensitive operations. the LT1533 greatly simplifies the task of eliminating supply noise by enabling the design of an inherently low noise switching regulator power supply. the LT1533 is a fixed frequency, current mode switching regulator with unique circuitry to control the voltage and current slew rates of the output switches. slew control capability provides much greater control over power sup- ply components that can create conducted and radiated electromagnetic interference. the current mode control provides excellent ac and dc line regulation and simplifies loop compensation. current mode control a switching cycle begins with an oscillator discharge pulse which resets the rs flip-flop, turning on one of the output drivers (refer to block diagram). the switch current is sensed across an internal resistor and the resulting volt- age is amplified and compared to the output of the error amplifier (v c pin). the driver is turned off once the output of the current sense amplifier exceeds the voltage on the v c pin. the toggle flip-flop ensures that the two output drivers are enabled on alternate clock cycles. internal slope compensation is provided to ensure stability under high duty cycle conditions. block diagra w + v c nfb fb r t c t sync gnd duty 1533 bd r vsl r csl oscillator t bk qb q ff sq ff r slew control internal v cc shdn v in pgnd col a col b output drivers + negative feedback amp + + g m error amp 1.25v 100k 50k + comp ldo regulator
7 LT1533 output regulation is obtained using the error amp to set the switch current trip point. the error amp is a transcon- ductance amplifier that integrates the difference between the feedback output voltage and an internal 1.25v refer- ence. the output of the error amp adjusts the switch current trip point to provide the required load current at the desired regulated output voltage. this method of controlling current rather than voltage provides faster input transient response, cycle by cycle current limiting for better output switch protection and greater ease in compensating the feedback loop. the v c pin serves three different purposes. it is used for loop compensation, current limit adjustment and soft starting. during normal operation the v c voltage will be between 0.2v and 1.33v. an external clamp may be used for lowering the current limit. a capacitor coupled to an external clamp can be used for soft starting. the negative voltage feedback amplifier allows for direct regulation of negative output voltages. the voltage on the nfb pin gets amplified by a gain of C 0.5 and driven onto the fb input, i.e., the nfb pin regulates to C 2.5v while the amplifier output internally drives the fb pin to 1.25v as in normal operation. the negative feedback amplifier input impedance is 100k (typ) referred to ground. slew control control of output voltage and current slew rates is done via two feedback loops. one loop controls the output switch collector voltage dv/dt and the other loop controls the emitter current di/dt. output slew control is achieved by comparing the currents generated by these two slewing events to currents created by external resistors r vsl and operatio u r csl . the two control loops are combined internally to provide a smooth transition from current slew control to voltage slew control. internal regulator most of the control circuitry operates from an internal 2.4v low dropout regulator that is powered from v in . the internal low dropout design allows v in to vary from 2.7v to 23v with virtually no change in device performance. when the part is put into shutdown, the internal regulator is turned off, leaving only a small (12 m a typ) current drain from v in . protection features there are three modes of protection in the LT1533. the first is overcurrent limit. this is achieved via the clamping action of the v c pin. the second is thermal shutdown that disables both output drivers and pulls the v c pin low in the event of excessive chip temperature. the third is under- voltage lockout that also disables both outputs and pulls the v c pin low whenever v in drops below 2.5v. 50% duty cycle mode since the LT1533 has dual out-of-phase outputs, it is ideal for driving push-pull transformers. for simple dc trans- former applications, the part can be forced into a 50% duty cycle mode using the duty pin. grounding the duty pin will override the internal control circuitry and force the outputs to switch with a 50% duty cycle at one-half the oscillator frequency. slew control also applies in the 50% duty cycle mode. applicatio n s i n for m atio n wu u u reducing emi from switching power supplies has tradi- tionally invoked fear in designers. many switchers are designed solely on efficiency and as such produce wave- forms filled with high frequency harmonics that then propagate through the rest of the power supply. the LT1533 provides control over two of the more impor- tant variables for controlling emi with switching inductive loads: switch voltage slew rate and switch current slew rate. the use of this part will reduce noise and emi over conventional switch mode controllers. because these variables are under control, a supply built with this part will exhibit far less tendency to create emi and less chance of wandering into problems during production.
8 LT1533 applicatio n s i n for m atio n wu u u it is beyond the scope of this data sheet to get into emi fundamentals. an70 contains much information concern- ing noise in switching regulators and should be consulted. oscillator frequency the oscillator determines the switching frequency and therefore the fundamental positioning of all harmonics. the use of good quality external components is important to ensure oscillator frequency stability. the oscillator is a sawtooth design. a current defined by external resistor r t is used to charge and discharge the capacitor c t . the discharge rate is approximately ten times the charge rate. by allowing the user to have control over both compo- nents, trimming of oscillator frequency can be more easily achieved. the external capacitance c t is chosen by: c t(nf) = 2180/[f osc(khz) ? r t(k w ) ] where f osc is the desired oscillator frequency in khz. for r t equal to 16.9k, this simplifies to: c t(nf) = 129/f osc(khz) , e.g., c t = 1.29nf for f osc = 100khz nominally r t should be 16.9k. since it sets up current, its temperature coefficient should be selected to compliment the capacitor. ideally, both should have low temperature coefficients. when the duty pin is high or floating, the outputs will be turned off during the discharge time of the oscillator. due to slew rate control, turning off the outputs does not produce immediate transitions. turn-off will require the current to ramp down and the switch voltage to ramp up. if the duty pin is grounded, then the outputs will turn on or off starting with the clock discharge. if the fb pin is below 0.4v the oscillator discharge time will increase, causing the oscillation frequency to decrease by approximately 6:1. this feature helps minimize power dissipation during start-up and short-circuit conditions. oscillator frequency is important for noise reduction in two ways: 1) the lower the oscillator frequency the lower the harmonics of waveforms are, making it easier to filter them, 2) the oscillator will control the placement of output frequency harmonics which can aid in specific problems where you might be trying to avoid a certain frequency bandwidth that is used for detection elsewhere. oscillator sync if a more precise frequency is desired (e.g., to accurately place harmonics) the oscillator can be synchronized to an external clock. set the rc timing components for an oscillator frequency 10% lower than the desired sync frequency. drive the sync pin with a square wave (with greater than 1.4v amplitude). the rising edge of the sync square wave will initiate clock discharge. the sync pulse should have a minimum pulse width of 0.5 m s. be careful in syncing to frequencies much different from the part since the internal oscillator charge slope deter- mines slope compensation. it would be possible to get into subharmonic oscillation if the sync doesnt allow for the charge cycle of the capacitor to initiate slope compensa- tion. in general, this will not be a problem until the sync frequency is greater than 1.5 times the oscillator free-run frequency. slew rate setting setting the voltage and current slew rates is easy. external resistors to ground on the r vsl and r csl pins determine the slew rates. determining what slew rate to use is more difficult. there are several ways to approach the problem. first, start by putting a 50k resistor pot with a 3.9k series resistance on each pin. in general, the next step will be to monitor the noise that you are concerned with. be careful with measurement technique (consult an70). keep probe ground leads very short. usually it will be desirable to keep the voltage and current slew resistors approximately the same. there are circum- stances where a better optimization can be found by adjusting each separately, but as these values are sepa- rated further, a loss of independence of control will occur. starting from the lowest resistor setting adjust the pots until the noise level meets your guidelines. note that
9 LT1533 slower slewing waveforms will dissipate more power so that efficiency will drop. you can also monitor this as you make your slew adjustment by measuring input and out- put voltage and current. it is possible to use a single slew setting resistor. in this case the r vsl and r csl pins are tied together. a resistor with a value of 2k to 34k (one half the individual resistors) can then be tied from these pins to ground. emitter inductance a small inductance in the power ground minimizes a potential dip in the output current falling edge that can occur under fast slewing, 25nh is usually sufficient. greater than 50nh may produce unwanted oscillations in the voltage output. the inductance can be created by wire or board trace with the equivalent of one inch of straight length. a spiral board trace will require less length. positive output voltage setting sensing of a positive output voltage is usually done using a resistor divider from the output to the fb pin. the positive input to the error amp is connected internally to a 1.25v bandgap reference. the fb pin will regulate to this voltage. internal regulator voltage (2.4v typ), output regulation may be disrupted. a series resistance with the feedback pin can eliminate this potential problem. negative output voltage setting negative output voltage can be sensed using the nfb pin. in this case regulation will occur when the nfb pin is at C 2.5v. the input bias current for the nfb is C25 m a (i nfb ) which needs to be accounted for in setting up the divider. referring to figure 2, r1 is chosen such that: rr v ra out 12 25 25 2 25 = - + ? ? ? ? ? . . m applicatio n s i n for m atio n wu u u fb pin 1533 f01 v out r2 r1 referring to figure 1, r1 is determined by: rr v out 12 125 1 =- ? ? ? ? . the fb bias current represents a small error and can usually be ignored for values of r1 || r2 up to 10k. one word of caution. sometimes a feedback zero is added to the control loop by placing a capacitor across r1 above. if the feedback capacitively pulls the fb pin above the figure 1 a suggested value for r2 is 2.5k. the nfb pin is normally left open if the fb pin is being used. dual polarity output voltage sensing certain applications may benefit from sensing both posi- tive and negative output voltages. when doing this each output voltage resistor divider is individually set as previ- ously described. when both fb and nfb pins are used, the LT1533 will act to prevent either output from going beyond its set output voltage. the highest output (lightest load) will dominate control of the regulator. this technique would prevent either output from going unregulated high at no load. however, this technique will also compromise output load regulation. shutdown if the shutdown pin is pulled low, the regulator will turn off. the supply current will be reduced to less than 20 m a. nfb pin i nfb 1533 f02 ? out r2 r1 figure 2
10 LT1533 applicatio n s i n for m atio n wu u u where d i is the ripple current in the switch, r csl and r vsl are the slew resistors and f osc is the oscillator frequency. power dissipation p d is the sum of these three terms. die junction temperature is then computed as: t j = t amb + (p d )( q ja ) where t amb is ambient temperature and q ja is the package thermal resistance. for the 16-pin so q ja is 100 c/w. for example, with f osc = 40khz, v in = 10v, 0.4a average current and 0.1a of ripple, the maximum duty cycle is 44%. assume slew resistors are both 17k and v sat is 0.26v, then: p d = 0.176w + 0.094w + 0.158w = 0.429w in an s16 package the die junction temperature would be 43 c above ambient. frequency compensation loop frequency compensation is accomplished by way of a series rc network on the output of the error amplifier (v c pin). referring to figure 3, the main pole is formed by capacitor c vc and the output impedance of the error amplifier (approximately 400k w ). the series resistor r vc creates a zero which improves loop stability and tran- sient response. a second capacitor c vc2 , typically one- tenth the size of the main compensation capacitor, is sometimes used to reduce the switching frequency ripple on the v c pin. v c pin ripple is caused by output voltage ripple attenuated by the output divider and multiplied by the error amplifier. without the second capacitor, v c pin ripple is: v vgr v c pin ripple ripple m vc out = ()( )()() 125 . where v ripple = output ripple (v p-p ) g m = error amplifier transconductance r vc = series resistor on v c pin v out = dc output voltage thermal considerations computing power dissipation for this ic requires careful attention to detail. reduced output slewing causes the part to dissipate more power than would occur with fast edges. however, much improvement in noise can be produced with modest decrease in supply efficiency. power dissipation is a function of topology, input voltage, switch current and slew rates. it is impractical to come up with an all-encompassing formula. it is therefore recom- mended that package temperature be measured in each application. the part has an internal thermal shutdown to prevent device destruction, but this should not replace careful thermal design. 1. dissipation due to input current: pvma i vin in =+ ? ? ? ? 11 60 where i is the average switch current. 2. dissipation due to the drivers saturation: p vsat = (v sat )(i)(dc max ) where v sat is the output saturation voltage which is approximately 0.1 + (0.4)(i), dc max is the maximum duty cycle. 3. dissipation due to output slew using approximations for slew rates: p vi i r iv v rf slew in csl in sat vsl osc = () + ? ? ? ? ? () ? ? ? () + () - ? ? ? ? ? () ? ? ? () ? ? ? ? ? ? ? ? ? ? () 2 2 9 2 2 9 4 33 10 4 220 10 d note if v sat and d i are small with respect to v in and i, then: p ir v r fvi slew csl in vsl osc in = ()( ) () ? ? ? + () () () ? ? ? ? ? ? ? ? ? ()()() 33 10 220 10 99
11 LT1533 applicatio n s i n for m atio n wu u u turns ratio of the transformer. the turns ratio must be large enough to ensure that the transformer can put out a voltage equal to the output voltage plus the diode under minimum input conditions. n vv dc v v out f max in min sw = + - () 2 () dc max is the maximum duty cycle of each driver with respect to the entire cycle which consists of two periods (q1a on and q1b on). so the effective duty cycle is 2 ? dc max . the controller, in general, determines maxi- mum duty cycle. a 44% maximum duty cycle is a guaran- teed value for this part. figure 3 v c pin 1533 f03 r vc 2k c vc 0.01 f c vc2 4.7nf to prevent irregular switching, v c pin ripple should be kept below 50mv p-p . worst-case v c pin ripple occurs at maximum output load current and will also be increased if poor quality (high esr) output capacitors are used. the addition of a 0.0047 m f capacitor on the v c pin reduces switching frequency ripple to only a few millivolts. a low value for r vc will also reduce v c pin ripple, but loop phase margin may be inadequate. magnetics design of magnetics is dependent on topology. the fol- lowing details the design of the magnetics for a push-pull converter. in this converter the transformer usually stores little energy. the following equations should be consid- ered as the starting point to building a prototype. figure 4 v in v sec t1 d s1 1 : n q1b q1a l o c o v out d s2 + 1533 f04 the following definitions will be used: v in = input supply voltage v sw = switch-on voltage v out = desired output voltage i out = output current f = oscillator frequency v f = forward drop of the rectifier duty cycle is the major defining equation for this topology. note that the output l and c basically filter the chopped voltage so duty cycle controls output voltage. n is the some common turns ratios v in v out n 5 10% 12 3.6 5 10% 15 4.4 5 10% 3.3 1.1 remember to add sufficient margin in the turns ratio to account for ir drops in the transformer windings, worst- case diode forward drop (v f ) and switch-on voltage (v sw ). there are a number of ways to choose the inductance value for l o . we suggest as a starting point that l o be selected such that the converter is continuous at i out(max) /4. if your minimum i out is higher than this, or you are operating at low currents such that the ic and components can handle higher peak currents, then use a higher number. continuous operation occurs when the current in the inductor never goes to zero. discontinuous operation occurs when the inductor current drops to zero before the start of the next cycle and can occur with small inductors and light loads. there is nothing inherently bad about discontinuous operation, however, the converter control and operation is somewhat different. the inductor is smaller for discontinuous operation but the peak currents in the switch, the transformer, the diodes, inductor and capacitor will be higher. but for low power situations these may not present a big constraint.
12 LT1533 applicatio n s i n for m atio n wu u u for continuous operation the inductor ripple current must be less than twice the output current. the worst case for this is at maximum input (lowest dc) but we will evaluate at nominal input since the i out /4 is somewhat arbitrary. note when both inputs are off, inductor current splits between outputs and the diode common goes to 0v. looking at the inductor current during off time, output ripple current is: d i out = 2 ? i out(min) i out(min) = i out(max) /4 l v dc if o out nom out = - () 12 d the inductance of the transformer primary should be such that l o , when reflected into the primary, dominates the input current. in other words, we want the magnetizing current of the transformer small with respect to the current going through the transformer to l o . in general, then, the inductance of the primary should be at least five times that of l o . this ensures that most of the power will be passed through the transformer to the load. it also increases the power capability of the converter and reduces the peak currents that the switch will see. l pri = 5 ? l o /n 2 if the magnetizing current is below 100ma, then a smaller l o can be used. with the value of l o set, the ripple in the inductor is: d i v dc lf out out o = - () 12 however, the peak inductor current is evaluated at maxi- mum load and maximum input voltage (minimum dc). ii i lmax out max out max =+ () () d 2 the magnetizing ripple current can be shown to be: d i vv nl f mag out f pri = + and the peak current in the switch is: i sw(peak) = n ? i lmax + d i mag this should be less than the 1a current limit. in the push-pull converter the maximum switch voltage will be 2 ? (v in C v sw ) plus a small amount (10%) for leakage spikes. because voltage is slew-controlled, the spikes will be less than normal. so, maximum switch voltage is: v sw(max) = 2 ? v in ? 1.1 this should be below the maximum rated switch voltage. so, given the turns ratio, primary inductance and current, the transformer can be designed. as an example: v in = 5v 10%, v out = 12v, i out(max) = 150ma, v sw = 0.5v, v f = 0.5v, f = 50khz, n = + () - () = 12 0 5 2 044 45 05 355 . . . . . round up so n = 3.6. for continuous operation at i out(min) = i out(max) /4, inductor ripple is: d i ma ma out == 2 150 4 75 the duty cycle for nominal input is: dc vv nv v l ma khz h nom out f in nom sw o min = + () - () = + () - () = = - () = 2 12 0 5 2365 05 38 6 12 1 2 38 6 75 50 730 . . . .% .% () () m off-the-shelf components can be used for this inductor. say we found an 800 m h inductor (coiltronics ctx200-1 for instance).
13 LT1533 applicatio n s i n for m atio n wu u u output ripple current at maximum input (dc = 34.7%) is: d i h khz ma out = - () = 12 1 2 34 7 800 50 92 .% m the maximum inductor current is: ima ma ma lmax =+= 150 92 2 196 primary inductance should be greater than: l h h pri == 5 800 36 309 2 . m m the magnetizing ripple current is approximately: d i h khz ma mag = + = 12 0 5 3 6 309 50 225 . . m peak switch current is: i sw(peak) = 3.6 ? 196ma + 225ma = 930ma which is less than the 1a maximum switch current. note that you can discern your magnetizing ripple by looking at the reflected inductance ripple and subtracting the switch current ripple. d i mag = n ? d i l C d i sw with knowledge of turns ratio and primary inductance along with volt/sec requirements (to prevent saturation) the transformer can be designed. transformers are available from coiltronics for some standard applications. figure 5 lists them. variations are available from coiltronics at 561-241-7876. also, see linear technologys application notes an19, an44 and an70 for further information about magnetics. capacitors correct choice of input and output capacitors can be very important to low noise switcher performance. push-pull topologies and other low noise topologies will in general have continuous currents which reduce the requirements figure 5. transformers for typical applications for capacitance. however, noise depends more on the esr of the capacitors. input capacitors must also withstand surges that occur during the switching of some types of loads. some solid tantalum capacitors can fail under these surge conditions. design note 95 offers more information but the following is a brief summary of capacitor types and attributes. aluminum electrolytic: low cost and higher voltage but in general dont use with this part because of high esr and poor high frequency performance. specialty polymer aluminum: panasonic has come out with their series cd capacitors. while they are only avail- able for voltages below 16v, they have very low esr and good surge capability. solid tantalum: small size and low impedance. typically available for voltages below 50v. possible problem with surge currents (avx tps line addresses this issue). os-con: lower impedance than aluminum but only avail- able for 25v or less. form factor may be a problem. sometimes their very low esr can cause loop stability problems. 2 3 primary a primary b section a section b = tied together * = high turns ratio version of ctx-02-13716-x1. accommodates low supply voltages or high dropout regulators 1533 f05 12 10 4 5 9 7 a 2 3 primary a primary b section a tie output common to this point section b 12 10 4 5 9 7 b nominal input voltage 5v 5v 5v 5v 5v 5v output power 1.5w 3.0w 1.5w 3.0w 1.5w 10w connection diagram a a b b a a coiltronics part number ctx02-13716-x1 ctx02-13665-x1 ctx02-13713-x1 ctx02-13664-x1 ctx02-13834-x3* ctx02-13949-x1 nominal output voltage after linear regulator 12v 12v 15v 15v 12v 12v
14 LT1533 applicatio n s i n for m atio n wu u u ceramic: generally used for high frequency and high voltage bypass. if all ceramic capacitors are used, they can have such a low esr as to cause loop stability problems. often they can resonate with their esl before esr be- comes effective. input capacitor the requirements for the input capacitor are less stringent for this part. input current ripple is lower because of the push-pull action and low noise features of the part. how- ever, the input capacitor should have low esr at high frequencies since this will be an important factor concern- ing how much conducted noise is created. values of input capacitor will typically be in the 1 m f to 22 m f range with esr under 0.3 w . the input capacitor can see a high surge current when a battery of high capacitance source is connected live. some solid tantalum capacitors can fail under this condi- tion. several manufacturers have developed a line of solid tantalum capacitors specially tested for surge capability (e.g., avx tps series). however, even these units may fail if the input voltage approaches the maximum voltage rating of the capacitor. avx recommends derating capaci- tor voltage by 2:1 for high surge applications. output filter capacitor output capacitors are usually chosen on the basis of esr since this will determine output ripple. typical required esr will be in the 0.05 w to 0.3 w range. the specific value for capacitance will depend on topol- ogy. a typical output capacitor is an avx type tps, 22 m f and 25v with a guaranteed esr less than 0.2 w . to further reduce esr, multiple output capacitors can be used in parallel. the value in microfarads is not particularly impor- tant. a small 22 m f tantalum capacitor will have high esr and higher output voltage ripple. table 1 shows some typical surface mount capacitors. table 1 size capacitor esr (max w ) e case avx tps, sprague 593d 0.1 to 0.3 avx taj 0.7 to 0.9 d case avx tps, sprague 593d 0.1 to 0.3 avx taj 0.9 to 2.0 panasonic cd 0.05 to 0.18 c case avx tps 0.2 (typ) avx taj 1.8 to 3.0 b case avx taj 2.5 to 10 switching diodes in general, switching diodes should be schottky diodes such as 1n5818 or mbr130 (1a/30v). low output current applications may use 1n4148 switching diodes. unregulated applications the LT1533 can be used to create a low noise dc transformer unregulated power supply. dc transformers are open-loop switching regulators where the output voltage is controlled by the turns ratio of the transformer. a dc transformer provides a low cost isolated supply. for such applications, the duty pin of the LT1533 should be grounded. this will force the outputs into a 50% on, 50% off mode. note that because of slew control there will be some variance from 50%. figure 6 shows a 5v to 12v dc transformer. one concern with this type of application is having both switch outputs transition at the same time. this can cause both primary side windings to have positive emf added to the winding, causing the current to run away. since this part controls slew rate this wont happen. it is possible to see slightly increased total current draw when both drivers are on, but this will be controlled and observable. since the outputs share a common sense resistor, the outputs will turn off when the total current in both exceeds the limit set by the v c pin. the fb pin should be dc biased between 0.7v and 1.2v to prevent frequency shifting from occurring. this also en- sures that the v c pin is set to its upper clamp, providing peak output current.
15 LT1533 applicatio n s i n for m atio n wu u u t he slew rate adjustment should be made by putting a 3.9k resistor in series with a 50k pot on the r vsl and r csl pins (or a 2k resistor in series with a 25k pot with both pins tied together). monitor output noise or other system signal while increasing the resistance until desired noise performance is reached. system efficiency can also be monitored. while this topology is not as quiet as a push-pull con- verter, it can provide a low cost, isolated power supply that has decreased noise relative to other solutions. more help an70 contains much information concerning LT1533 applications and measurement of noise and should be consulted. a 5v to 12v demo board is also available (dc173). an19 and an29 also have general knowledge concerning switching regulators. our application depart- ment is always ready to lend a helping hand. typical applicatio n s u figure 6. 5v to 12v dc transformer 18k 10k 43k LT1533 gnd r t v c nfb fb 5v 22 m f 10v 25nh* t1** l2 100 h l1 100 h 68k 4k to 68k v in duty shdn sync 1 1 3.3 3.3 1533 f06 3300pf 12v 80ma ?2v 80ma 1n5819 4 lt1121cs8 lt1175cs8 22 m f 35v 22 m f 35v r4 150k 150k 2.2 m f 25v 2.2 m f 25v 332k 324k 81 32 1, 2, 7, 8 3 4 5 14 2 16 15 12 13 7 5v 8 10 5 6 9 4 11 3 bat85 bat85 c t shdn * bead or pcb trace ** coiltronics ctx02 13716-x1 l1, l2: coilcraft dt1608c-104 col a pgnd r csl r vsl col b
16 LT1533 typical applicatio n s u low noise 5v to C12v forward push-pull converter. output noise is below 100 m v. noise performance is identical to positive output version. see an70 for details shdn duty sync col a col b pgnd r vsl r csl nfb fb LT1533 gnd v in 5v 14 2 15 16 13 12 8 11 3 4 5 6 10 3300pf 18k 0.01 f 15k l1 4.7 f 15k + t1 1n4148 1n4148 l2 100 h l3 100 h 2.4k 1% 7 9 9.6k 1% 47 f + + 47 f l1: 22nh inductor. coilcraft b-07t typical, trace inductane or bead l2, l3: coiltronics ctx100-3 t1: coiltronics ctx02-13665-x1 ?2v 1533 ta03 c t r t v c optional for lowest ripple () shdn duty sync col a col b pgnd r vsl r csl fb nfb LT1533 gnd v in 2.7v to 4v (3 nicd batteries) 14 2 15 16 13 12 7 11 3 4 5 6 10 3300pf 18k 0.01 f 15k l2 4.7 f 15k + t1 l1 100 h l3 100 h 3.48k 1% 8 9 21.5k 1% 47 f + 47 f 9v 1n4148 1n4148 l1, l3: coiltronics ctx100-3 l2: 22nh trace inductance, ferrite bead or inductor coilcraft b-07t typical t1: ctx02-13665-x1 1533 ta04 + c t r t v c optional for lowest ripple () electronic equivalent of 9v battery operates from three nicd cells. output noise is below 100 m v. see an70 for details
17 LT1533 typical applicatio n s u a 50v output low noise regulator. cascoded bipolar transistors accommodate 60v transformer swings, permitting 24v (20v in to 30v in ) powered operation. see an70 for details hysteretic loop lowers quiescent current to 100 m a while maintaining low output noise. see an70 for details shdn duty sync col a col b pgnd r vsl r csl fb nfb LT1533 gnd v in 2.7v to 4v (3 ni-cd batteries) 14 2 15 16 13 12 7 11 3 4 5 6 10 3300pf 18k 0.01 f 10pf 5v 15k l2 4.7 f 15k + t1 l1 100 h l3 100 h 8 9 21.5k 1% 2.32k 1% 150k 1.18v v z internal to ltc1440 47 f + 47 f 12v 1n4148 1n4148 l1, l3: coiltronics ctx100-3 l2: 22nh trace inductance, ferrite bead or inductor coilcraft b-07t typical t1: ctx02-13665-x1 1553 ta05 + c t r t v c optional for lowest ripple () + c1 ltc1440 shdn duty sync col a col b pgnd r vsl r csl nfb LT1533 gnd v in 24v (20v to 30v) 14 2 15 16 13 12 11 3 4 5 6 10 3300pf 18k 0.01 f 15k 3.3k 3.3k l2 1 f q3 1n4148 68 15k + t1 l1 100 h 50v 100 h 8 7 9 97.6k 1% + + 47 f 47 f mur110 mur110 l1: coiltronics ctx100-3 l2: 22nh trace inductance, ferrite bead or inductor coilcraft b-07t typical q1, q2: zetex ztx-853 q3: 2n2222a t1: ctx02-13665-x1 1553 ta06 47 f c t r t v c optional for lowest ripple () 360 + 360 68 2.49k 1% 10k 1n752 5.6v q1 0.003 f 0.003 f q2 fb
18 LT1533 a 10w low noise 5v to 12v converter. q1-q2 provide 5a output capacity while preserving LT1533s voltage current slew control. efficiency is 68%. higher input voltages minimize follower loss, boosting efficiency above 71%. see an70 for details typical applicatio n s u shdn duty sync col a col b pgnd r vsl r csl fb nfb LT1533 gnd v in 5v 14 2 15 16 13 12 7 11 3 4 5 6 10 1500pf 18k 0.01 f 10k 2.49k 1% l2 fb2 fb1 4.7 f 10k + t1 l1 300 h 12v 5a l3 33 h 8 9 21.5k 1% + + 100 f 100 f 1n5817 1n5817 1n4148 1n4148 1553 ta07 c t r t v c optional for lowest ripple () 330 330 0.05 0.05 4.7 f + q2 q1 0.003 f 680 l1: coiltronics ctx300-4 l2: 22nh trace inductance, ferrite bead or inductor coilcraft b-07t typical l3: coiltronics ctx33-4 q1, q2: motorola d45c1 t1: coiltronics ctx02-13949-x1 fb1, fb2: ferronics ferrite bead 21-110j
19 LT1533 information furnished by linear technology corporation is believed to be accurate and reliable. however, no responsibility is assumed for its use. linear technology corporation makes no represen- tation that the interconnection of its circuits as described herein will not infringe on existing patent rights. dimensions in inches (millimeters) unless otherwise noted. package descriptio n u s package 16-lead plastic small outline (narrow 0.150) (ltc dwg # 05-08-1610) 0.016 ?0.050 0.406 ?1.270 0.010 ?0.020 (0.254 ?0.508) 45 0 ?8 typ 0.008 ?0.010 (0.203 ?0.254) 1 2 3 4 5 6 7 8 0.150 ?0.157** (3.810 ?3.988) 16 15 14 13 0.386 ?0.394* (9.804 ?10.008) 0.228 ?0.244 (5.791 ?6.197) 12 11 10 9 s16 0695 0.053 ?0.069 (1.346 ?1.752) 0.014 ?0.019 (0.355 ?0.483) 0.004 ?0.010 (0.101 ?0.254) 0.050 (1.270) typ dimension does not include mold flash. mold flash shall not exceed 0.006" (0.152mm) per side dimension does not include interlead flash. interlead flash shall not exceed 0.010" (0.254mm) per side * **
20 LT1533 1533f lt/tp 0598 5k ? printed in usa ? linear technology corporation 1997 typical applicatio n u related parts part number description comments lt1129 700ma micropower low dropout regulator 0.4v dropout voltage, reverse battery protection lt1175 500ma negative low dropout micropower regulator positive or negative shutdown logic lt1377 1mhz high efficiency 1.5a switching regulator high frequency, small inductor lt1425 isolated flyback switching regulator excellent regulation without transformer third winding ltc ? 1436 high efficiency synchronous switching regulator adaptive power tm mode, phase locked loop lt1534 ultralow noise 2a switching regulator ultralow noise regulator for boost topologies adaptive power is a trademark of linear technology corporation. 10w off-line power supply passes fcc emission requirements without filter components r csl r vsl LT1533 fb col a col b d s d s 1v q5 irf840 q6 irf840 1n5818 1n5818 c t 3.9k shdn v in v c r t pgnd 43k 3300pf l2 22nh 4v 0.48v 1 m f 8.2k 4v 12v 470k 75k 0.8 w bat-85 22k hv 12v 12v 10m 1k 360k 10k 12v 12v 12v 15k 0.002 m f + 1 m f + 220 m f + 15 m f 1 m f + 10 m f + 4.7 m f 1n759a 12v + 330 w 240k 2.5v r top col lt1431 fgnd 1533 ta08 ref sgnd r mid 0.15 m f + c1 1/2 lm393 + c2 1/2 lm393 power limit current limit 3k l1 = coiltronics up-4 l2 = coilcraft b07t npn = 2n3904 unless otherwise noted pnp = 2n3906 t1 = coiltronics ctx02-13978-x3 1v 510 w 0.48v 470 w q1 mpsa42 q2 5k 0.5w 100k hv + 1k q3 q4 1k 360k t1 10k 470 w 4n28 12v 0.002 m f hv 1.6k 1w 1.6k 1w 0.001 m f 250v 0.001 m f 250v l1 10 m h 5v 2a q7 q8 hv 0.1 m f ac line 1n4005 100 m f 400v + = 20cjq045(i.r.) unless otherwise noted = 1n4148 = ac (hot) return = output common danger!! high voltage!! screened area contains lethal high voltages! use caution in construction and testing! linear technology corporation 1630 mccarthy blvd., milpitas, ca 95035-7417 (408) 432-1900 l fax: (408) 434-0507 l www.linear-tech.com

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